Tag Archives: mixers
T3 mixers are the highest dynamic range mixer available. They are also handbuilt parts, subject to unit to unit and lot to lot variability. In this blog post we attempt to quantify that variability. Our sample is 10 T3-08LQP mixers from 5 different date codes. All the date codes are separated by at least a month, totaling nearly two years. Therefore, the variation you see in the plots below accurately represent the variation a designer could expect across two years in the life of their product. Of course there are always outliers, but the following represents typical performance variation.
Marki is bringing advanced mixer designs to a broader market with four new models of GaAs Schottky diode double balanced mixers covering S and K band applications. These designs combine the legendary mixer design expertise of Marki Microwave with the repeatability and economies of scale intrinsic in the MMIC production method.
Sometimes you need a mixer; sometimes you need an IQ mixer. How do you know which one to buy? Before answering this question, I recommend reading the Mixer Basics Primer to get a good understanding of the fundamentals of mixers, the blog post ‘IQ, Image Reject, and Single Sideband-Mixers’ for and introduction to these mixers, and ‘How to think about IQ mixers’ if you want a deeper physical understanding of the mechanisms of IQ, image reject (IR), and single sideband (SSB) mixers.
The oldest question in mixer tech support is probably “what happens when I drive the mixer with X dBm LO?”, where X is some number lower than what we recommend. In general, and particularly in the past, we have avoided this question. Our recommendation was and is to never underdrive a mixer. The reason for this is that a mixer with insufficient LO drive does not act as a switching device, but as a square law device. If the LO does not turn on the diodes then the physics of mixer operation change completely, and all of our carefully laid design work is thrown out the window.
Indeed, when you underdrive a mixer the conversion loss is not the only thing that changes. All of the specs change, and in unpredictable ways. The LO side in particular responds weirdly, because a lot of deficiencies and inefficiencies on the LO side are exposed when it is underdriven and concealed under normal operation. In this post we will show all the bad things that can happen when you don’t supply an adequate LO drive to the mixer, and then leave it to you as the user to decide what LO drive to design with.
In our last post we showed the physical basis for how mixers are used as phase detectors, concluding by showing that IQ mixers make ideal phase detectors due to their ability to unambiguously identify the relative phase between two signals at any power level. In this post we examine the opposite: how to use mixers as phase modulators. It seems like you should be able to use them in exactly the opposite way, which is to apply a DC voltage to get a linear phase shift. Unfortunately, it’s not that simple.
Note: as with mixers as phase detectors, we as the manufacturers are not the best experts, but our users are. In this case I would recommend Kratos General Microwave, whose application notes I used in preparation of this blog post.
Why phase modulators?
Before examining how to get a phase modulator, let’s look at why you might need one. The main applications are communications and electronic warfare.
Communications: Phase modulation (mathematically identical to frequency modulation) has been used since very early in radio communications, due to FM communications having constant amplitude, better spectral/power efficiency, and convenience. The most common way of understanding phase modulation is with binary phase shift keying (BPSK), or quadrature phase shift keying (QPSK) if both orthogonal components are used. All modern communication systems use these techniques, so they have been written about very extensively, and we will assume that you are familiar with them.
Electronic Warfare: Here it gets interesting. If you have a phase modulator in a jammer, you can trick an enemy radar system into thinking that your plane/boat/tank is not where it actually is. You do this by listening to their radar pulses and responding with frequency shifted radar pulses, making it appear that you are moving at a different speed. This is the classic decoy technique. Modern jammer systems employ much more advanced, exotic, and classified schemes than this that I hope I never have the classification level to learn about. The principles, however, are the same.
Double Balanced Mixers as Phase Modulators
Let us start by running a phase detector in reverse. Instead of a DC output, lets input a DC signal to the IF port and a CW signal to the LO port and see what comes out of the RF port. If the device is reciprocal, then a small DC voltage/current should induce a small phase change, and a larger voltage should create a larger phase change. Here is what happens:
So this is nothing like what we expected. Why? I don’t really know. Somewhere the hand waving ‘superposition’ argument I gave in the last blog post breaks down, and something is not reciprocal. The above behavior makes sense from what we know about double balanced mixers. Namely that with no voltage applied, the LO-RF isolation prevents any signal from passing through. As the DC voltage is applied, it breaks down the symmetry of the diode quad, reducing the isolation and allowing more of a signal to pass through, although without a phase change. Now here is what happens when we apply a negative voltage:
So at least with a negative voltage you can get a 180° phase flip. So a double balanced mixer does give you phase modulation, but only between two different options. This makes it suitable as a BPSK modulator (emphasis on the word binary), but not for much else.
IQ Mixers as Phase Modulators
IQ mixers worked great as phase detectors, will they work great as phase modulators? Only one way to find out:
This picture is a little confusing, but the idea is that it works out mostly okay. As with the double balanced mixer, you see no signal pass through with 0 voltage applied. If you manipulate the DC voltage applied to the I and Q port, then you will see the phase rotate around the complete circle as expected. When you only have one voltage applied you will be almost at 0°, 90°, 180°, or 270°. There are some phase errors, I would imagine due to the non-ideality of the components, but this can probably be accounted for. There is also a non-uniformity to the amplitude due to the isolation being degraded differently in the different mixers.
So what is going on here, how can you achieve an arbitrary phase with an IQ mixer when you can only get two phases with the double balanced mixers that make them up? This is the following trigonometric identity:
a cos(x) + b sin(x) = R cos(x-theta)
where R2=a2 + b2 and tan(theta) = b/a. This means that by simply changing the values of the input signals (in this case by modulating the isolation of the mixers) you can achieve any phase within the range of the tangent function (-90° to 90°), and then by flipping the value to negative you can achieve any value in the phase circle.
Is this a good idea?
Just because you can do something doesn’t mean you should, you can drive a car with your feet if you want to but that doesn’t make it a good idea.
The short answer is yes and no. Using an IQ mixer is an easy way to achieve an arbitrary phase modulator, definitely useful if you are in the lab. A few problems though:
- The isolation change with applied DC voltage is non-linear, and the whole structure has to be carefully characterized to achieve repeatable results.
- All non-idealities are frequency dependent, so this characterization has to take place at each point in the system. Further, if you are modulating a broadband or multitone signal you won’t be able to correct for the system errors at all frequencies.
- The insertion loss when used in this way will look very little like the insertion loss shown in the datasheet for the IQ mixer. For example, here is the MLIQ-0416 datasheet conversion loss:
Nice and flat across the band around 8 dB. Now here is the same mixer as a phase modulator:
This starts out strong, but then falls off at higher frequencies. This is because the second plot is of an insertion loss instead of a conversion loss. The difference is subtle but important. In the first case the LO is only used to turn on the diodes, so the losses that it takes passing through the LO quad hybrid don’t matter very much. In the second case, the LO is the signal, so any loss that it takes shows up as an insertion loss.
Those are the main problems, and if these can be overcome, then you may be in business.
Single Sideband Mixer as a Phase/Frequency Modulator
There is another way to use an IQ mixer as a phase or frequency modulator, and this involves creating quadrature CW signals into the IF port of the mixer, effectively using the IF port as the LO, and varying the frequency of this LO signal to change the frequency offset. We will examine this, and other use cases, in our next blog post on using IQ mixers as single sideband upconverters and image reject downconverters.
Some of the most common questions we receive here are about using mixers as phase detectors. We previously discussed this topic in the post, “DC Offset and Mixers as Microwave Phase Detectors”. In this post we will go into much further depth about the physical mechanisms by which mixers act as phase detectors, and what is important for engineers trying to accomplish this in the lab. First a warning though: we’re just showing experimental results here. The real experts in phase detectors, phase noise, and all things related to phase are the people that do this every day at Holzworth Instrumentation.
Double Balanced Mixers as Phase Detectors
Much has been written about how double balanced mixers work as phase detectors (for example, see this article from Watkins Johnson about the subject). As with most circuit topics the descriptions in the literature are based in math rather than physical principles, so we’ll now consider the physical mechanisms in play when a double balanced mixer is used as a phase detector. Let’s look at what happens when we apply two in phase (frequency matched) voltage signals to an ideal double balanced ring mixer1:
The phases correspond to the phase of each signal as it appears at the diodes. Only two are show, but take my word that superposition works here and every other in between state produces the same effect. For current to flow, two conditions must be met. First there must be a voltage differential across the diode. Second is that the diode has to be pointed in the correct direction. The red arrows indicate where those conditions are met and current will flow. As you can see, when the signals are in-phase current will flow into the IF balun, creating a positive DC voltage at the IF port. Now out of phase signals:
The situation is similar for out of phase signals, except that current is always pulled out of the IF balun, thus creating a negative DC voltage at the IF port. For quadrature signals, there is equal current flowing both into and out of the IF balun. This means that no net DC current is created, no net voltage is apparent. The IF port is essentially always a DC virtual ground. This is the physical basis for why a double balanced mixer will show no DC voltage for two signals in quadrature2.
With these principles understood, let’s go in the lab and see what happens when we actually apply these signals to double balanced mixers. First we create two signals and use an oscilloscope to verify that they are in quadrature:
At 10 GHz the period is 100 ps, so 25 ps out of phase is in quadrature. Now we apply these voltages to the input of a ML1-0220LS mixer, and what do we find? Nonzero voltage! In fact, here is what the DC output voltage (taken with a terminated bias tee, this is very important) looks like a function of phase between the two input signals:
Now we would expect this to be a peak at 0°, and the minimum to be at 180°. What is going on? This is a phenomenon that is documented by Stephan Kurtz in the previously referenced WJ app note. In modern double balanced mixers the RF and LO baluns are not identical. In fact, they are not even close. One side is built as a magic tee, where the IF is removed, and the other side has a return to ground on it. Even though the LO and RF baluns traditionally cover identical frequency bands, there is no reason that they need to. They can be completely different! This means that they most likely have a different electrical transmission length and phase delay, which is why the peak of the voltage curve is not quite at 0°. Another effect highlighted in this app note is that there is a voltage offset that shifts the entire curve up (or down). As we detailed in our first post, the excellent balance and isolation of the ML1-0220 minimizes this DC offset and makes it not noticeable for this plot.
1 Note that the necessary DC current return to ground path is necessary but not illustrated for clarity.
2 It is easy to imagine how to extend these principles to the situation where the signals are not at their peak or zero values, and similarly to phases that are not either perfectly in phase or out of phase. While superposition does not strictly work in a nonlinear system such as this, the results one would expect from superposition are maintained qualitatively.
IQ Mixers as Phase Detectors
Now we can calculate the phase of the signal. Excellent. However, there are two ambiguities that we need to clear up. Since this is a sine wave instead of a sawtooth wave, there is some ambiguity about the phase. The same output voltage could be two different phases, except for the max and min. This is fine if you are doing phase noise testing, where you put the two signals in quadrature and just look at any voltage that comes out. For actually detecting the phase between two signals though, it isn’t enough information. The second ambiguity is that we need to know the max and min voltage levels, as well as the DC offset, to determine the phase. Since the DC offset in Microlithic mixers is small we can ignore this, but we still have a problem if the incoming signals change power at all.
How do we resolve this? One way is to use two mixers as phase detectors and deliberately introduce a phase shift between the two inputs (RF and LO). 180° is no good, because the phase ambiguity remains, so a balun is out. A length of line changes phase with frequency, so that is out too. The other broadband phase shifting options we have are a Schiffman phase shifter or a quadrature hybrid. The quad hybrid is much more common and easy to build3, so what would a structure with a quadrature hybrid introduced on one side look like?
That’s right: the structure is exactly an IQ mixer. Since I and Q are in quadrature, it is easy to calculate the phase between the two signals as
after making a small correction to scale the I and Q values by their peak output level and DC offset4. Let’s look at the same plot of voltage vs. phase for the IQ mixer, along with the calculated phase:
As expected, the calculated phase is almost linear with input phase after the correction factors. This is a significant improvement over the double balanced mixer, since we don’t need to know the input power levels and there is never any phase ambiguity. But how close is the IQ mixer to invariant with input power? When the two signals are at 0/90/180/270 degrees to each other, there is obviously very little variation in calculated phase with power since one of the voltages doesn’t change. If we pick a phase in the middle (135°)5, this is what it looks like:
As you can see the power levels the agreement to 135° is excellent. As we increase to higher power levels, one of the mixers compresses sooner than the other mixer, and the phase is thrown off. Up to 0 dBm, however, the agreement with the real phase is excellent. This does not address what happens when one of the signals is significantly higher than the other one, nor with double balanced mixers when you are just trying to detect phase changes, where high powers are desirable to increase sensitivity.
Now that we have examined the physical mechanisms of how mixers work as phase detectors, we can do the reverse and see how they work as phase modulators. This is the subject we will tackle in our next post, “All About Mixers as Phase Modulators”.
3Quad hybrids are easier than Schiffman phase shifters, but still ridiculously difficult to build broadband. You don’t have to trust me, you can try yourself, and then buy ours when it takes you 6 months.
4You also have to convert to a -180 to +180 phase range, or 0 to 360, or whatever. Arctangent only gives you values from -90 to +90, so you have to use the sign of the signals to figure out where exactly you are.
5How do we know the phase is 45 degrees? Because we put the signals in quadrature (which is the same at any power levels), and then moved them 12.5 ps on the oscilloscope, equivalent to 45° at 10 GHz.
Spectral regrowth is a big deal for you. In order for the wireless revolution to continue apace, enabling you to watch funny cat videos faster in more crowded environments, spectral regrowth must be conquered wherever it occurs. Spectral regrowth is what occurs when a broadband or spread spectrum signal intermodulates with itself, creating deterministic products that look like noise, limiting the signal to noise ratio of the signal. According to Shannon’s theorem this limits the information capacity of the signal, and thus your video takes longer to load (for some reason this always happens at the worst time).
Spectral regrowth comes from a handful of sources. It can come from mixers, but in installed communication systems it tends to come from the power amplifiers at the transmitter and the connections to the antenna itself (called ‘passive intermodulation’ or PIM). It is made much worse by using higher power and by denser concentrations of signals. Both of these factors are increasingly common as data capacity is increased. This is why highly linear amplifiers and PIM are both big buzzwords in the mobile communications world right now.
What is not always talked about is that ‘spectral regrowth’ in the mobile communication world is the same as ‘two-tone intermodulation distortion’ or ‘IP3’ in the microwave world. Two tone modulation distortion is what causes spectral regrowth, just summed over all of the frequencies involved in the signal. This can be seen by moving from two tone testing to three tone testing in a standard double balanced mixer, the Marki M1-0212SA:
As you can see, it is a complete mess! This is considered very good for a double balanced mixer, and it is better than you would get from any GaAs mixer on the market. This is with 0 dBm output signals and a 25 dBm LO drive (square wave, this will matter in a minute). Note that we use 0 dBm output power as the reference instead of the input power. As we have mentioned before, this is what you really care about as a designer (how much range do I have at the final stage in my receiver) and accounts for the variation in insertion loss between mixers. You can cheat by adding loss to a mixer to improve the input IP3, but you can’t cheat on the output IP3.
This output spectrum is obviously unsuitable for operating with, nonetheless testing a high performance system. The testing system must have at least 20 dB more dynamic range than the system itself, and usually much more, so getting rid of these obnoxious intermodulation products is critical. This was the state of the art for many years, then Watkins Johnson came out with their termination insensitive mixer, the M8TH (still on sale from Ma/com and still considered the standard by some). Here is the M8TH output spectra:
Much better! The intermodulation products have been suppressed significantly, although they are still significantly limiting the dynamic range to 45 dB. Recently there has been much talk of the FET mixer. At Marki we have not been that impressed with the FET mixer, because of the narrow bandwidth over which it works. Combined with the intrinsic poor isolation available in the FET circuit, this limits the performance below what we like to see. Nonetheless people get excited about it, so here is the narrow bandwidth, poor isolation PE4140 FET quad mixer we built two and three tone results:
Good improvement above the termination insensitive mixer, now the dynamic range is up to 50 dB. At least I guess it is cheap and good for communication applications, but with the narrow bandwidth and isolation problems it isn’t well suited as a test and measurement mixer. Now let’s look at the T3. As a true commutating mixer it is in its element with a 25 dBm square drive. Here are the results:
And this is also the mixer with the best spurious suppression, and isolations and conversion loss as good as any mixer available. All of this across a 2000:1 bandwidth ratio. It isn’t fair to say that the T3 is the Ferrari of mixers. It’s more like the T3 is a fighter jet racing a car. When it comes to choosing a mixer for your test system, there really is no choice but the T3.
There are many ways to think about IQ modulation, and all of them rely on math. This is because ‘quadrature’ modulation is a mathematical construct, a way of thinking about how time domain signals can be manipulated more than a physical reality. In this blog post I will describe how I think of IQ modulation, which is as the cancellation of a signal through two 90° phase shifts that create a 180° phase shift, which is the negative of the original signal. The negative and positive versions of the signals cancel, resulting in suppression of the other signal. This is identical to the math that governs image cancellation in image reject and single sideband mixers, the only difference is that one of the 90° phase shifts occurs at the transmitter in an IQ scheme, while they are both at the receiver in the image reject/single sideband scheme.
The easiest way to see this is with a combination of a trigonometric derivation and graphics. I will try to make the math as straightforward as possible, since I don’t speak math well.
We’ll use the following trigonometric identities:
Consider a single sided downconversion imagining the mixer as a perfect frequency multiplier. The output at the RF will be given (ignoring the 2πt terms) by
The same math works for an upconversion:
It is irrelevant whether we use an in phase (cosine) or out of phase (sine) LO for a double sided downconversion on a single tone, other than a phase shift of the output (90° from the input and 180° between the two products) for a downconversion:
Or an upconversion:
Now consider following the double sided upconversion with a double sided downconversion. We’ll multiply the original output by an in phase LO
Where we use the identities given above. Clearly it can be seen that the desired IF frequency is present along with the undesired 2*LO terms. However, if we attempt to perform a downconversion with an out of phase LO, the following results:
As you can see from the math and the diagram, the two sidebands compete at the downconverted sideband, canceling each other out. For this reason a double sided upconversion followed by a double sided downconversion is not recommended. If the phase of the LO is set correctly the signal will be reconstructed with twice the amplitude of a single sided downconversion, but if it isn’t phased correctly the two sidebands will cancel each other.
This same phenomenon can easily be shown to occur if a sine wave LO is used as the upconverting LO and a cosine is used as the downconverting LO. This raises an interesting possibility, however. If a phase coherent LO is available, then we can upconvert one signal into a sideband and downconvert it with the same phase LO (with some gain). We can also upconvert a separate signal using a 90° out of phase LO and transmit across the same medium, downconverting it with the same LO but again 90° out of phase. The sidebands of the signal will cancel each other out for the out of phase signal while adding constructively for the in-phase signal. This is called quadrature modulation, and is the basis for such modern signaling techniques as quadrature amplitude modulation (QAM), which is the how all modern wireless communications systems operate.
Here is what the math looks like (it gets a little messy, you can just look at the conclusions):
Combined upconverted signal, with signals added:
Now multiply by an in-phase LO:
After low pass filtering we recover:
The in-phase components add constructively, while the quadrature components add destructively. It can be similarly shown for a quadrature (sine wave) downconverting LO, we will only recover the b(t) signal. Graphically this appears like this:
At this point we can see everything that we need to make an IQ modulator: 2 matched mixers, a device to separate the LO into in-phase and quadrature signals (called a quadrature hybrid coupler) and a device to add the two signals together (an in-phase power combiner).
Before we move on from the IQ modulator, consider what would happen if we eliminated one of the sidebands after the upconversion and try to downconvert:
After low pass filtering, this becomes:
That is, both versions of the signal are present without cancellation or suppression, and neither can be recovered without the information present in the second sideband. There are many more advanced modulation techniques using DSP that may offer quadrature information transmission in a single sideband, but this cannot be achieved using conventional analog components.
Every day we work on high linearity mixers: high IP3, high P1dB, and high spurious suppression. Every once in a while we get a request for a high IP2 mixer. This is much more rare than complaints about IP3 or spurs. Lets see why.
To start understanding IP2 in mixers, lets look at intermodulation products in amplifiers. Start by imagining a single tone into a single non-ideal amplifiers, with a nonlinearity.
A single input tone is amplified to a larger tone, as desired. Due to the nonlinearity in the amplifier, higher order tones are created. These are only created at integer multiples of the input tone, since the system is time invariant. These decrease in power as the frequency increases, and they are generally referred to as second, third, Nth order distortion products.
A new problem arises if we put two tones into the same amplifier:
This diagram only shows the second harmonics, the second order intermodulation, and the third order intermodulation in the relevant bands. As you can see, the second harmonics (at 2f1 and 2f2) are easily filtered out as they are at a very high frequency, unless you are in a very broadband multioctave system. The second order intermodulation (at f1 + f2) is in between the two harmonic distortion products, so it is also easily filtered except in broadband systems. While these are drawn as the same power level, they are not necessarily the same power in practice. The third order harmonics (at 2f1-f2, f1+f2-f1, f1+f2-f2, and 2f2-f1) are all famously in band. Two of the tones are directly on top of the desired received tones, and therefore they cannot be filtered even with an infinitely narrow and steep filter.
Now to mixers. The addition of the time varying LO increases the complexity dramatically. All of the previous effects will be present, and also will all be present at the LO crossed frequencies. For the moment lets ignore everything but the isolations, fundamentals, and second order distortions (both single and multitone).
In this case, there are two types of second order products: direct and converted. The direct products are at the same frequency as in an amplifier, and the converted second order product is between the 2IF x 1LO products. Again these are drawn as the same power level, but they may or may not be identical powers.
When will these products matter? The converted products will appear in the passband in a broadband system (with a low IF) where they cannot be filtered out. Conversely the direct second order products will matter in a different kind of broadband system, with a high IF. Specifically when fIF = 1/3 fLO, the converted signal and the second harmonic of the IF, and the second order distortion product, will all be at 2/3 fLO. These can be a problem, but usually no worse and closely affiliated with the 2IF x 0 LO spur.
So in both cases we see that while the second order distortion exists, it is always close to a high power spur that also must be dealt with in the frequency plan.
Now lets consider a downconversion:
Once again the converted second order shows up, this time in between the two 2LO – 2RF spurs that usually wreak havoc on downconversion systems. Once again the same frequency plan is needed to eliminate it. The direct second order term, however, is at f2-f1, which becomes a significant problem when the IF frequency is similar in magnitude to the separation between the two tones. In this case the direct second order tone would lie directly over one of the tones.
The converted two tone second order intermodulation product will be an issue in the same circumstance as the 2x-2 spur is a problem, namely when you have a low IF. If the IF is at DC (direct downconversion) then the second order intermodulation will cause significant distortion at DC. This is why the most common reference to IP2 in the literature is for the mitigation of it in direct downconversion receivers.
One thing that is not IP2, but is sometimes referred to as IP2, is the half-IF spur. This occurs when a signal (at Frf) is downconverted to a low frequency, near baseband, and there is a jamming signal at a frequency (Fj) roughly halfway between the RF and LO frequencies. The downconverted jamming signal can be filtered out by the IF filter, along with all other unwanted signals. However, the jammer signal creates a high power 2 LO x 2 RF spur, however, that will show up at or near the desired signal, and there is no physical way to filter it out.
Fortunately, in either a double or triple balanced mixer structure the 2×2 spur will be well canceled by both the LO and RF baluns, resulting in excellent suppression when well balanced baluns are used. For example, the ML1-0218ISM offers a downconversion 2×2 suppression of 58 dBc with an input of -10 dBm. The T3 circuit can offer even better suppression, since the proprietary T3 circuit will both prevent and suppress these spurs. Therefore the T3-18 offers a superior 64 dBc suppression of the 2×2 spur with the same -10 dBm input.
However, this is not a two tone IP2 problem. It is simply a second order distortion product. So I shouldn’t take these authors too hard to task. As Joel Dunsmore cautions in his book, Introduction to Microwave Measurements:
There is sometimes confusion in the use of the term second-order intercept; while it is most commonly used to refer to the second harmonic content, in some cases, it has also been used to refer to the two-tone second-order intercept, which is a distortion product that occurs at the sum of the two tones. Most properly, one should always use the term two-tone SOI if one is to distinguish from the more common harmonic SOI.
And that is the final point of this post; when you are talking about IP2, you always need to be specific about what you mean.
In making the datasheets for the first Microlithic frequency doubler (MLD-1640), it occurred to us that not enough has been made about the difference between isolation and suppression.
In mixers and amplifiers, some parameters are expressed relative to the input powers, while some are expressed in terms of the output power, with the conversion loss or gain calibrated out. This includes third order intercept point (IP3), which can be expressed as either input IP3 (IIP3), or output IP3 (OIP3). In general it is better to use OIP3 for mixers, since what really affects the dynamic range of a system is the amplitude difference between the output signal and output spur, expressed in dB relative to the output signal or carrier (dBc). This is illustrated in the table below, where the difference between the T3 and competing mixers is even greater when the superior conversion loss of the T3 is considered.
|T3-05||33 dBm||6.5 dB||26.5 dBm|
|Imitator 1||25 dBm||10.7 dB||14.3 dBm|
|Imitator 2||30 dBm||9 dB||21 dBm|
Note that it is better to use IIP3 in amps, for the opposite reason, namely that you want to give the amp credit for it’s gain. So in parts with gain the appropriate measure is IIP3, while in parts with a loss the appropriate spec is OIP3.
When the same logic is applied to spurious products in mixers and multipliers, the input referred value in dB is called isolation, while the output referred value in dBc is called suppression. Suppression is the preferable number to use, because it expresses the important value to the system. The isolation can always be improved by increasing the conversion loss of the mixer or multiplier, but this is obviously undesirable. There are, however, some issues using suppression.
The first comes with mixers. In all mixers we express the spurious output of the LO in terms of isolation, since it is dependent on the input LO power. Since the LO power does not change the conversion loss referenced to the input, this means that the suppression can vary by several dB with different LO drive levels.
The second complication is that the input signal, converted signal, and spurious tone are all at different frequencies. For example, when using a doubler with an 8-20 GHz input range, the output doubled frequency is 16-40 GHz, and the undesired tripled frequency is 24-60 GHz. This means that the isolation curves look like this:
While the third harmonic suppression looks like this:
The curves look slightly different. The suppression is more stretched out, and distorted by the curve of the conversion loss of the doubler. This is the result of the suppression calculation, which can basically be thought of as a 1:1 mapping of the isolation through the conversion loss.