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T3 mixers are the highest dynamic range mixer available. They are also handbuilt parts, subject to unit to unit and lot to lot variability. In this blog post we attempt to quantify that variability. Our sample is 10 T3-08LQP mixers from 5 different date codes. All the date codes are separated by at least a month, totaling nearly two years. Therefore, the variation you see in the plots below accurately represent the variation a designer could expect across two years in the life of their product. Of course there are always outliers, but the following represents typical performance variation.
At Marki, as in much of the microwave industry, we tend to focus on frequency domain measurements. This includes S-parameters (insertion loss, return loss, isolation, rejection), parameters that can be calculated from S-parameters (amplitude balance, phase balance, common mode rejection ratio, directivity), and related parameters (group delay flatness, differential group delay). These are the parameters that are most interesting for applications such as radar, narrow band communications, and electronic warfare.
In the signal integrity industry, the people that design the high speed signaling hardware that powers backplanes on major computing systems powering the internet, among other applications, time domain is king. These engineers generally care about one thing: the data throughput that a system is capable of. Components are evaluated on the basis of whether they can successfully transmit the data. For this reason the primary metrics are eye diagrams, error vector magnitude plots, and bit error ratio measurements.
Each approach has its benefits and drawbacks, and it is important for a well rounded engineer to master both domains. Think of time domain as an aerial view and frequency domain as a microscope. The drawback of frequency domain measurements are that they can be too specific, and you can miss the forest for the trees. You know the 3 dB rolloff is at 7 GHz, and the return loss has a troubling hump at 3 GHz, but will that sink the ship or not?
In time domain measurements, you immediately see the big picture, but if anything is wrong it is difficult to diagnose. For example, a ringing in the eye could be from many sources, and there may be no way to isolate the source without resorting to frequency domain measurements. Here are a few eye distortions, and what the potential sources might be. First, here is what your basic eye diagram looks like, as plotted by microwave office:
It is perfectly square and with minimal ringing because I selected many sample points. If you select fewer points, you get an eye that looks more realistic, like this:
Which is the same way that it looks
after you filter it with a filter that looks like this
except with some ringing due to Gibb’s phenomenon, a result of the limited high frequency content in the signal. Ringing is among the most common phenomena experienced in a lab, but it is generally harmless, since it does not introduce noise near the sampling point. In fact, a signal can be filtered very aggressively without affecting the sampling point:
as shown in this example with a 500 MHz filter (1/2 the datarate). On the other hand, high pass filtering will cause baseline wander, which shows up in eye diagrams as a split in the different levels of eyebrows that is dependent on the coding scheme of the PRBS or data sequence:
In the 128 bit sequence, there are discrete levels that result from each bit sequence. If this bit sequence repeated then it would always look like that. The eyebrow is not necessarily from random noise, as it appears, but from deterministic artifacts of a given bit sequence. This is why coding like 8B/10B and 64B/66B can be used on lower bandwidth hardware: the low frequency components are not distorted because they are not present.
What we’ve covered so far is only half the story. There’s still phase/group delay and timing variations, which are usually much more important than amplitude considerations, but that will have to wait for another post.
As a relative newcomer to microwave and RF, there are certain things that the industry takes for granted that I find very weird when I first encounter them. It took me a long time to really understand the concepts of isolation and directivity, for example, and why do 2.4 mm connectors look just like SMA connectors when they don’t fit together? Couldn’t we color code them or something?
S-parameters are one of these things. I know from working with our datasheets and test data that the S parameters represent power loss. For example, and insertion loss of 3 dB corresponds to a power loss of ~50%. This is because converting 10 dB to linear units, you use the formula 10^(-3dB/10) = .50118723362727, the point is that 3 dB is approximately a 50% power loss, though not exact. There is no fundamental reason this is so, it just conveniently happens to be about the same.
According to our microwave bible, “Microwave Engineering” by David Pozar, the scattering matrix [S] is defined according to the forward and reverse voltage waves as [V-] = [S][V+]. A specific element of the matrix is defined as a formula, but the formula states roughly that “Sij is found by driving port j with an incident wave of voltage V+j, and measuring the reflected wave amplitude, V-i, coming out of port i. This is the definition that is used in all of the books and papers that I have seen using scattering parameters.
Here is the tricky part, that everyone in the microwave industry takes for granted, and has been hazy for me for a while. If the S-parameters are defined in terms of voltage, than a 3 dB reduction should mean a 50% reduction in voltage, equivalent to a 75% reduction in power. A 3 dB loss (or gain) however never means a 75% reduction in power, or a 50% reduction in voltage. This is because:
S parameters in linear units always refer to the amplitude (voltage or current), while S parameters in logarithmic (dB) units always refer to power.
There, I said it. It is one of those things that is usually gets lost or taken for granted going from the classroom to the lab. It’s like amplitude balance: even though it is called amplitude balance, it refers to the the difference in power between two outputs, and it is quoted in dB. So when calculating between linear and logarithmic S parameters, you have to use 10^(X dB/20), while for power it is 10^(X dB/10). It’s weird, and you just have to get used to it.
I hope this clears things up for some poor students out there. Good luck on your finals and happy late spring break!
I was thinking about the difference between power dividers, baluns, and couplers, and realized that they could all be thought of as power splitters. The characteristics that make them different are the relationship between the outputs in terms of amplitude, phase, and attenuation between outputs. Here is a brief chart that explains them all:
When you master phase, you become like a God, capable of performing wonders that mere mortals can only dream of. Wonders like making laser beams (using phase engineered quarter-wave reflectors), communicate tremendous information great distances through thin air (all modern communication formats use both amplitude and phase), and create amazing products (balanced amplifiers, balanced mixers, phased array antennas, Mach Zender modulators, the list never ends).
BUT…phase is the hardest thing to understand in microwaves, RF, and photonics. It is hard to measure, hard to visualize, and makes some very confusing homework problems that kept me in the late night coffeeshops of Champaign-Urbana well past my bedtime.
In this post we will make a dent in the universe of phase understanding by clarifying the difference between phase and group delay, and in the process explain why you can’t match phase with variable line lengths. When you buy a phase shifter, it is sometimes what I would call a real phase shifter, and sometimes what I would refer to as a ‘group delay shifter’. The trombone type variable delay lines (we like the ones from sage) are actually variable time delay elements, and not phase shifters.
A group delay (or time) shift is easy to understand: it is how long the pulse (or wave) takes to arrive at your measuring receiver. Differential delay is therefore the difference in how long it takes for two pulses or waves to arrive. In passive components it is just the distance divided by the speed of light (or whatever your wave is) at your frequency in your material.
Phase is much more difficult. It is the integral of group delay over frequency (plus an offset), or differently the group delay is the derivative of the phase vs. frequency. This is why filters can be used as time delays; the edges of the filter have significant phase variation that leads to significant group delay variations over a narrow bandwidth (this is called Kramers-Kronig relation).
A variable length delay line, therefore, can only change the phase by changing the group delay. But by changing the group delay, you are changing the integral (slope) of the phase vs. frequency. This means that the phase change will be different at different frequencies. This is very different than what you get from a quadrature hybrid coupler, or a balun, where the phase shift is constant across frequencies. The difference is shown below. First is a plot of the phase difference between the two outputs of a BAL-0520 Balun (180°), a QH-0226 quad hybrid (90°), a coupler plus two 37.5° Schiffman phase shifters we developed as a custom (165°), a PD-0220 wilkinson power divider (0°), and a PD-0220 with an extra .570″ adapter on one side (variable).
As you can see, the phase is flat across the bandwidth of the device for everything except the PD-0220 with the extra delay line (adapter). This has a rapidly changing phase across frequencies. If we take the derivative of this we should get the group delay, but instead I measured the differential group delay with the PNA-X.
Here you can see that the differential group delay between outputs for each of the devices is 0, except for the power divider with the adapter, which has a flat constant group delay (ignore the big hump, I think that is from the calculation the PNA is doing with the phase flip).
So what is the lesson? You can phase match two outputs using a variable delay line, but only at a single frequency. Otherwise you have to do it with a coupler, a balun, a Schiffman, or some other true variable phase circuit.
We frequently receive inquiries about the moisture sensitivity of our surface mount components.Technically, all of our components are moisture sensitivity level 1 (MSL1) according to the IPC/JEDEC’s J-STD-20, meaning that they can be mounted and reflowed an unlimited amount of time after they are removed from the packaging. This is not the complete story however.
Because of the unique, high frequency construction of our surface mount devices, the lid provided for them is only a dust cover, epoxied on. For this reason it is not sealed against aqueous solutions, and special precautions need to be made.
All sensitive elements inside of the units are sealed independently, and the circuits themselves are not moisture sensitive. However, during the aqueous wash process the solution will penetrate the components and therefore will require some time to dry.
For this reason, Marki Microwave suggests that all surface mount components be subject to a vacuum bake at less than 120° C for one hour.
The important element in this is the vacuum, which will cause the solution inside to evaporate and evacuate the dust cover. A lower temperature or lower vacuum can be used, but it must be subject to the process for a longer time. If the part is tested within 24 hours after an aqueous wash, then the performance will be affected by the solution still inside the dust cover.
Microwave designs are pushing towards an all surface mount future, due to reduced assembly complexity and cost and reduced system size. Marki Microwave is pushing this technology by offering many of our designs in surface mount packages. The main question for a microwave surface mount package is the transition: how do you bring the signals from the board into the component, and back? The component must be on a substrate of some sort, and a straight wire up the side of the substrate will not generally be 50 ohms. The trick is to maintain a broadband 50 ohm impedance through the transition. Marki uses three different techniques to achieve this:
Image of T3A-EZ package
EZ eyelet – The EZ package uses a Marki proprietary technology we call an ‘eyelet’. This is a 50 ohm transition built into a metal carrier to allow a transition to a taller base height for the substrate. A suspended substrate is then attached to the carrier, allowing for surface mount suspended substrate mixers. This is the best method for transitioning to a suspended substrate part available.
Product Lines: Mixers (M1, M3, M4), Amplifiers, Doublers
Carrier Material: Tin/Lead(85/15) plated Brass
Max Recommended Frequency: 20 GHz
Suspended Substrate Compatible: Yes
Lead-Free Option: Because of the construction of the EZ eyelet, it cannot be built with lead free solder. For this reason the M1, M2, and M3 mixers are not available as lead-free surface mount units.
Image of T3-CQ package
CQ and CQG Castellation Via – A castellation via is a method of creating a transition by making a plated through hole in a circuit and cutting the through hole in half. Unlike in an EZ eyelet, it does not have a ground that transitions with it, so the transition appears as an inductive line. Further, it is limited to certain materials that are capable of supporting the construction. It is however very robust and is a visible transition, making it easier to inspect the solder fillets.
Product Lines: T3 Mixers
Carrier Material: FR4
Max Recommended Frequency: 16 GHz
Suspended Substrate Compatible: No
Lead-Free Option: Yes (CQG)
Bias Tee SM Package
Plated Through Hole – This is a standard way to achieve surface mount transitions. It can be performed with a wider variety of materials, including thinner and lower dielectric materials.Because it is used with thinner materials, it is typically only associated with smaller parts, to keep the board flat and prevent it from warping during assembly. Because it is used with thinner materials the transition can appear less inductive, thus performing to higher frequencies.
Product Lines: Bias Tees, Power Dividers, Diplexers
Carrier Material: PTFE
Max Recommended Frequency: 35 GHz
Suspended Substrate Compatible: No
Lead-Free Option: Yes (SMG)
We are happy to announce that you can now search our product catalog on Google’s shopping for suppliers at http://www.google.com/shopping/suppliers.
We are excited to be selected as a pilot company for this innovative new way to shop that we expect to be extremely useful for our customers in the future.
Space is a really tough environment. Electronic parts must suffer massive shock and vibration during launch and often see wide temperature swings as satellites are heated by the sun then slip behind the earth into very cold outer space. Most space parts must tolerate a vacuum and heat must be managed carefully. Radiation hardening is critical as a spacecraft is bombarded by a merciless sea of high-energy particles. Parts must be clean and outgassing limited, to ensure that camera lenses are not clouded and there is little tolerance for repairs. Designing and building parts for space requires tenacity and a commitment to process… from managing ESD, testing, and training to handling analyses correctly and efficiently.
Procurement specifications for space are usually written for the requirement at hand. A scientific mission with a tiny budget will rely on the expertise of the manufacturer to assist in defining the technical details of the part and testing. A commercial satellite will have more detailed specification often defined at the satellite level, rather than for the particular assembly required as part of the satellite. Costs can mount as program managers sift through technical data and handle all contingencies. And missing a deadline is not an option.
Standardizing specification around existing MIL standards and specifying hardware already designed for space can reduce costs, improve lead times and overall, improve the quality of life. A paper that I delivered at PTTI in Reston, VA provides more details on the benefits of this approach. Read Paper
Thank you Christopher Marki for the wonderful engineering and personal insights that you shared in the Microwave Journal. We read them, passed them around, and felt at one with you as an engineer in a small business who cares about the technology (the stuff) and your family, co-workers and customers (the people).
At Wenzel Associates we are a close group of engineers, technologists and manufacturers, people who also love the stuff and the people. We make very low noise oscillators, frequency based systems and synthesizer to about 16 GHz. There are lots of custom modules shipped from our Austin, Texas facility, some in complicated nickel-plated aluminum boxes with Spira shields to reduce cross-coupled spurs, some that slide into rugged racks mounted in a fighter jet or mount on the cold plate of a satellite ranging system.
We hope to contribute bits of useful information to RFblogger, life’s lessons perhaps, things relevant to the RF field. I am Liz Ronchetti, President and co-founder of Wenzel Associates with my husband, Charles Wenzel, our CEO and the brilliant one in the family. John Richardson is our Head of Engineering at Wenzel and I bet we will see posts from him in the future. My BSEE degree is from Worcester Polytechnic Institute in frosty Worcester Massachusetts and I am grateful to be writing from warm and sunny Austin, Texas.
So what’s all this about 217Plus?
We recently ran into a source control drawing from a prime contractor that called out 217Plus, rather than MIL-HDBK-217F for its MTBF calculation. After a bit of research and a call to a very nice QA engineer at Lockheed who forwarded me to another, here is what I learned.
“Everyone knows that MIL-HDBK-217F is outdated,” having last been updated in 1995. Lockheed (among others) reviewed suppliers, looked at processes and histories and decided if an improved number was appropriate, sometimes for a single part, sometimes for a complete line. They applied a methodology on the numbers, making judgments and surveys and supplementing them with more data, always applying the same criteria. They sent their methodology to the Navy to verify that they were good with it. The Navy saw that there was a mismatch with what happens in the field and the 217F predictions, and that the new predictions gave results that were actually closer to what happens. In many of cases this approach was preferred by the Navy because it was more accurate and also kept needless costs down.
MIL-HDBK-217F defines the “failure rate” by calculating a rate for each subcomponent. The rate is assigned by part type. For example, the formulas needed to calculate failure rate for a ceramic capacitor (Fixed, ceramic, general purpose CK, CKR) differ from the formulas for an FET (Transistor, High Frequency, GaAs FET). Formulas for each part type’s specific reliability effects are detailed and multiplied together for a final “Failures/10e6 Hours”. In the case of the ceramic capacitors the factors are:
λb base failure rate => ambient and max rated temperature, stress (ratio of operating to rated voltage)
πCV capacitance factor => uses capacitance in pF
πQ quality factor => part quality type , S, R, …, MIL-C-11015 non-established reliability, and lower
πE environment factor => ground based GB, airborne inhabited cargo AIC, etc.
This modeling system, by its own admission, is limited. The information in a released standard can only be as up to date as the data at time of publication. The writers recognize that “Electronic technology is noted for its dynamic nature”, and write that reliability will certainly vary by the differences in system application, operational scenarios and even in the definition of failure.
It seems that, from lots of collected data and analysis, new standards have evolved, probably not as suddenly or as easily as it seems. Some companies, in conjunction with the DOD and the Reliability Information Analysis Center (RIAC), have built software around these standards. PRISM® software, originally released in 1999, is available through System Reliability Center (SRC). In 2006 the upgrade 217Plus was released by the RIAC and is available through SRC. Relex, from Parametric Technology Corporation, is the software that we use for our MTBF calculations, and it also supports PRISM® and the 217Plus upgrade. There may be other programs available.
Eventually, MIL-HDBK-217 will be updated to Rev G; it has had an initial release for review and comments, but is still a work in progress.
Why is this important
This is significant to RF designers and manufacturers, any of us who have scoured data sheets and poured through websites looking for a good MIL-style varactor. MIL parts are expensive. They are less available than they were and we are finding more MIL parts have become obsolete. What were our favorite parts may be no more because better more reliable parts are available and have replaced them. Many of these new parts are characterized in 217Plus.
So new designs can be cheaper and have a shorter development time; we are all grateful for this. Now, new reliability information is readily available, accessible and easy to use. The creation of 217Plus also suggests that we are getting better at producing high quality high reliability parts as a group. This means safer air travel, better radar systems and lower cost mobile radios.
Of course, we will use caution in applying 217Plus. More research is needed to learn the applicability of the new data and find out who is currently using it. There may be cases were the new numbers are not approved, such as in space work, or for some mission critical designs that were qualified a long time ago and any changes means the addition of risk. But it is very welcome.
Wenzel Associates, Inc.