## How to Measure Mixer IP3 and Identify Potential Error Sources

Mixer linearity is continuously and permanently a critical problem faced in RF system design. The nonlinear action of all physically realizable RF mixers propagates throughout signal chains generating undesired, unfilterable output harmonics, multitone intermodulation, and nonrecoverable nonlinear signal distortion. For example, nonlinear mixing action can cause undesired output harmonics (i.e., spurs) such as the $2f_{RF} \times 2f_{LO}$ or the $2f_{RF} \times f_{LO}$ spur instead of the desired $f_{RF} \times f_{LO}$ converted signal. In multitone applications, such as data transmission and radar tracking, mixing heavily exacerbates the problem of spectral purity by not only introducing a second set of unwanted, unfilterable output harmonics but by also introducing multitone intermodulation distortion (IMD). IP3, or the 3rd order intercept point (TOI), is the figure of merit by which industry judges the linearity of all active, power-consuming RF components and their ability to maintain core linear assumptions about circuits. The mixer is no exception.

A power calibrated vector network analyzer and an external driving synthesizer is the standard tool to do IP3 measurements. While VNAs have their own errors, such as limited RF input power and poor 2nd input harmonic suppressions, it is the most convenient and time efficient way to measure IP3 over broadband frequency sweeps. Spectrum analyzers, although highly susceptible to user error, provide a secondary way of doing comparable IP3 measurements. Both measurement on a spectrum analyzer and a VNA should agree. This document serves as a tool for debugging, developing, and utilizing the spectrum analyzer, it’s supporting synthesizers, and other RF paraphernalia for mixer IP3 measurements.

IP3 and IMD Explained

The most important thing to understand about the IP3 specification is that, like a mixer’s spurious suppression, it is a measure of your in-band spur-free dynamic range, and how linear your output signal is. IP3 is not a directly measured mixer parameter like conversion loss, return loss, or isolation. IP3 is a figure of merit of device linearity based on the relationship between a 2-tone analytic signal and its resulting self-intermodulation terms.

To calculate the input referred IP3 (IIP3) we use:

$IIP3 (dBm) = P_{Input} + \dfrac{P_{Fundamental}-P_{IM3}}{2}$

To calculate the output referred IP3 (OIP3) we use:

$OIP3 (dBm) = P_{Fundamental} - \dfrac{P_{Fundamental} - P_{IM3}}{2}$

$OIP3 (dBm) = IIP3 + Conversion Loss$

For the case where a mixer is used as a downconverter with a low side LO, $P_{Input}$ is the average power of the analytic 2-tone RF input signal. $P_{Fund}$ is the average power of the down converted 2-tone RF input to the frequencies $f_{RF_1}-f_{LO}$ and $f_{RF_2}-f_{LO}$. $P_{IMD3}$ is the average power of the spurious output that appears at $(2f_{RF_1}-f_{RF_2})-f_{LO}$ and $(2f_{RF_2}-f_{RF_1})-f_{LO}$. $P_{IMD3}$ is a mixing product of the 2-tone RF input and mixer generated second order intermodulation products to the $(2f_{RF_1}-f_{RF_2})-f_{LO}$ and $(2f_{RF_2}-f_{RF_1})-f_{LO}$ frequencies. For an upconverter, the principal is the same.

1. Depiction of where the 2nd and 3rd order IMD products fall in a mixer output spectrum based on a 2-tone RF input signal. Frequency plan based on a mixer used in a down conversion of $f_1$ and $f_2$ using a highside LO.

IIP3 is input referred IP3 and OIP3 is output referred. This can be measured in either an up-conversion or down-conversion and the result of which should be roughly equivalent for the same or similar frequency plan (RF, LO, and IF frequency bands and power levels). To visualize what OIP3 means, see figure 2 for what could typically be seen on a spectrum analyzer for a downconverted 2-tone signal.

2. Example output spectrum of an IP3 measurement. Output will include unwanted harmonics such as those close to the carrier.

First, there is the need to clarify the difference between IMD products and spurious products of a mixer. IMD tones are generated from multiple, close input fundamental frequencies, $f_{RF_1}$  or $f_1$ and $f_{RF_2}$ or $f_2$. They are not generated from harmonic mixing of the RF/IF signal and the LO signal. An example IMD product is the $(2f_{RF_1}-f_{RF_2}) - f_LO$ harmonic. Spurious products, such as the $1f_{LO}-3f_{RF}$ spur, are the result of the unwanted harmonic mixing of the RF/IF signal and the LO signal.

Similarly, for a mixer’s IP2, you’re not interested in measuring the power of the $f_{LO} - 2f_{RF}$ spur, you’re interested in the $1f_{LO}-(f_{RF_1}-f_{RF_2})$ or $1f_{LO}-(f_{RF_1}+f_{RF_2})$ spur.

3. DUT test frequency plan for a 2-tone RF input. Mixer tested as a down converter from 6GHz RF to 300MHz IF.

Testing Part I: Quick and Dirty

The frequency plan above is the frequency plan of the running example of this document. The example measurement in this document will be done as a down conversion with the MM1-0312HS (unless otherwise specified) from 6GHz +/- 0.5MHz RF to 299MHz +/-0.5MHz IF with a low side fixed 5.701GHz LO. For convention, RF refers to the high frequency small signal input that will be mixed down to the low frequency IF output. The RF signal is input into port 1, LO into port 3, and IF is output from port 2 (configuration B).

4. Mixer input/output signal configuration. RF signal is input into port 1, LO signal into port 3, and the IF signal is pulled from port 2.

First, we generate a combined 2-tone input signal with 2 synthesizers and any power divider (i.e., power combiner); for instance, the PD-0010 resistive power divider.

$f_1$ is the 6.0005GHz tone and $f_2$ is the 5.9995GHz tone. Both tones have +0dBm power, verified on both a filtered power meter and our Rohde and Schwarz FSW85 spectrum analyzer. It’s important to understand we’re trying to only test our mixer so we must always power calibrate to the plane of the mixer’s RF port. To avoid measurement error, we must calibrate to only the fundamental tone. Using an unfiltered power meter to do this measurement is slightly erroneous because power meters measure the total output power of a signal. This includes all harmonics in addition to the fundamental tone power.

5. Quick and dirty mixer IP3 measurement. Measurement may or may not be accurate with this setup.

We repeat the power calibration step for the LO source set at 5.701GHz, +15dBm. It is extremely important that sufficient LO power is used to fully turn on a mixer or else risk unpredictable results. Mixers are heavily LO power dependent, if LO power is varied, the IP3 measurement result can and will vary widely. If insufficient LO power is delivered, the conversion loss suffers heavily and will provide erroneous IP3 results.

6. IF output spectrum of the MM1-0312HS with the block diagram setup from figure 5.

After calibrating, the mixer is connected as shown in figure 6. When we calculate the IP3 value, we typically average both down converted tones and both IMD spurs. So, remembering that the data displayed on the screen of the spectrum analyzer is the mixer’s output, we can first calculate the mixer’s OIP3 and then back calculate the mixer’s IIP3 because we know the RF input power.

$OIP3 = \frac{(f_{Out_1}+f_{Out_2})}{2}+\frac{\frac{(f_{Out_1}+f_{Out_2})}{2}+\frac{(f_{IMD_1}+f_{IMD_2})}{2}}{2}$latex OIP3 =\frac{(-8.55+-8.28)}{2}+\frac{\frac{(-8.55+-8.28)}{2}-\frac{(-54.02+-53.93)}{2}}{2}=+14.4dBm$$IIP3 = OIP3 - Conversion Loss = 14.4-\frac{(-8.55+-8.28)}{2}= +23.9 dBm$ For a “quick and dirty” measurement the result is not bad. It relatively matches the value displayed on the datasheet. This was a straight forward measurement where we did not consider non-idealities in the measurement, and implicitly trusted our test equipment to report the actual IP3 value. As a rule of thumb, if the mixer’s IP3 is below +25dBm, this is a sufficient setup with modern test equipment. To provide more accurate data however, we must account for the non-idealities. Error Source #1 To account for all of the non-idealities in the measure we look mainly at the small signal RF input for the down conversion measurement. In figure 8, we see that the “quick and dirty” measurement’s spectrum is fairly clean by casual standards; the IMD3 spurs are not stronger than -80dBm. This establishes part of the IP3 measurement floor. This floor isn’t good enough for the most accurate mixer IP3 results. To get the best results, we need to decrease the IMD3 tone power of the input. Getting an ideal result of no input intermodulation products is not possible, although they can be greatly suppressed. 7. Spectrum of the 2-tone RF input into the MM1-0312HS with the block diagram from figure 5. The 2nd harmonic inputs at $2f_1$ and $2f_1$ are very low power (or should be). The $2f_1$ and $2f_2$ will mix down to $(2f_{RF_ 1} - f_{RF_2})- f_{LO}$ and $(2f_{RF_2} - f_{RF_1}) - f_{LO}$ and should be accounted for; a passive diode mixer will respond to any signal of any power. If an RF amplifier is used, the 2nd harmonic should be filtered off. 8. Diplexers added after the signal sources from figure 4. Lowpass port of the diplexer passes the signal. Input $2f_1$ and $2f_2$ terms terminated in highpass port with a 50Ω termination. Lowpass filter the 2nd harmonic of $f_1$ and $f_2$to eliminate a low power source of intermodulation within the mixer. This is done with the lowpass port of an applicable diplexer. Unwanted high frequency signals will not be reflected but instead be sunk into the highpass port and its 50Ω termination. 9. Spectrum of the 2-tone RF input into the MM1-0312HS with the block diagram from figure 8. There is a marginal gain in the 2-tone RF input IP3 as a result of the lowpass filters. Not surprisingly this results in no change in the mixer’s IP3 measurement. 10. IF output spectrum of the MM1-0312HS with the block diagram setup from figure 8. Error Source #2 Another major, but less obvious, source of error is the cross talk between the two RF input synthesizers. All synthesizers have a phase locked loop (PLL) to lock the output frequency to an internal (or external) reference frequency. When a reverse signal is present, either from a mismatched load reflecting power back, or from cross-talk introduced from an external signal source, the phase detector in the PLL will respond. Even if an RF amplifier is at the output of the PLL the amplifier reverse isolation is finite and will leak power into the phase detector. 11. Example phased locked loop showing a possible reverse signal path. The feedback path of the PLL will account for anymismatched reflection. If instead, a 2nd RF input synthesizer were to be the reverse signal, then the phase detector block of the PLL will produce its own harmonics from unwanted intermodulation due to the non-linear action of the phase detector (i.e., mixer). To reduce the unwanted reverse signal, we can do 2 things. First, we provide isolation through the power divider used to minimize the cross talk between the 2 RF input synthesizers. Second, we can provide additional isolation on each synthesizer arm through the use of attenuators or isolators. 12. RF input diagram using power divider that provides isolation. If instead of the PD-0010 resistive power divider we were to use a power divider that provides isolation, like the PD-0R413 Wilkinson power divider, the cross-talk should be reduced. Power traveling from one output port of the power divider to the other will be attenuated by the power divider’s isolation value. 13. Spectrum of the 2-tone RF input into the MM1-0312HS with the block diagram from figure 12 using the PD-0R413 power divider. Swapping in the PD-0R413 Wilkinson power divider reduces the 2 tone RF input’s IMD3 terms by 4dB. Using an even higher isolation power divider like the PBR-0006 should provide a slightly better result. 14. Spectrum of the 2-tone RF input into the MM1-0312HS with the block diagram from figure 12 using the PBR-0006 power divider. Incrementally increasing the power divider isolation marginally improves the measured output IP3 of the 2-tone RF input signal. While the 3rd order tones decrease, the higher order intermodulation products may change. Mixer IP3 measurement with the PD-0R413: 15. IF output spectrum of the MM1-0312HS with the block diagram setup from figure 12 using the PD-0R413 power divider. Mixer IP3 measurement with the PBR-0006: 16. IF output spectrum of the MM1-0312HS with the block diagram setup from figure 12 using the PBR-0006 power divider. The mixer’s measured IP3 is effectively the same. So we continue the example using the PBR-0006 because of the higher isolations. To improve the isolation 10dB attenuators are added to the circuit as seen in figure 18. The lowpass filters used in front of the synthesizers are DPX-9516s. They are reflective filters near the crossover point of the diplexer. Outside of the DC-9.5GHz lowpass filter passband, high frequency signals will reflect back towards their source. 17. 10dB attenuators added between the diplexer and power divider to improve the $f_1 to f_2$ isolation by 20dB. Possible signal paths shown in blue and red. What is of interest for the reflective filters is the device under test (DUT) mixer’s finite LO-RF isolation and the mixer’s internal harmonics generated from the fundamental LO. The LO signal can, does, and will leak through from the LO port to the RF port and output into the RF input synthesizers. By adding 10dB attenuators between the lowpass filters and power divider, the LO fundamental tone and LO harmonic spurs’ leak through is reduced by 10dB. Any LO harmonic reflections will be reduced by 20dB before remixing in the mixer. e.g., For a typical Marki GaAs MMIC double balanced mixer, the LO to RF isolation will be approximately 40dB. So an input of a+15dBm fundamental into the LO port will give -25dBm at the RF port that could remix if reflected. If the 2nd LO harmonic is strong enough, that can also remix. Unless the LO frequency is low, the higher order harmonics will be rejected (reflected) by the RF balun of the mixer. Error Source #3 Because the LO harmonics can be an issue, along with any passive intermodulation products, the power divider’s combined output should be filtered. 18. RF input block diagram from figure 18 with a diplexer presented to the mixer. Instead of using another a filter in front of the mixer’s RF port, a diplexer (not duplexer) is a good choice. A diplexer is wideband enough to provide a path for the 2nd and 3rd order LO to RF harmonics coming out of the RF port to be terminated in a 50Ω load. The diplexer used must be carefully chosen to avoid the LO harmonics from reflecting back into the mixer. Otherwise, the power divider is a better interface to present for impedance matching considerations. This provides a 1dB improvement in the IMD3 spur power that is input into the mixer. 19. Spectrum of the 2-tone RF input into the MM1-0312HS with the block diagram from figure 18. 20dB internal attenuation in the spectrum analyzer. There is no change to the measurement of the mixer’s IP3 from the addition of the 3rd diplexer in front of the mixer’s RF port. 20. IF output spectrum of the MM1-0312HS with the block diagram setup from figure 18. Error Source #4 All spectrum analyzers have a front-end mixer prior to their IF envelope detector circuitry; that mixer limits the dynamic range of the system. To make sure that the spectrum analyzer’s mixer does not generate its own harmonic content and disturb the measurement, the power into the spectrum analyzer must be attenuated to prevent overloading the receiver. Secondarily, we want to suppress the front-end mixer from re-mixing higher order products. If we check against the spectrum analyzer datasheet, our Rohde and Schwarz FSW85 has a typical IIP3 of +30dBm with a minimum of +22dBm for an input frequency of 100MHz to 1GHz. For our input powers (-8dBm on both fundamental IF tones) into the spectrum analyzer that floor is approximately -70dBm if we back calculate with our standard IP3 formula. So the IMD3 spur can be down ~-70dBm (with no internal attenuation) before the spectrum analyzer will hit its dynamic range limit. This is the second floor of the measurement that can fundamentally limit the measured dynamic range of the DUT. You must have a better part than the DUT within the test equipment to measure the DUT. As a precaution, we can filter the input of the spectrum analyzer to prevent high order harmonics from mixing within the spectrum analyzer. 21. IF output spectrum of the MM1-0312HS with the block diagram setup from figure 18. Using a DPX-0R5 to filter the mixer IF output, we see that the above measurement was not impacted or changed when we correct the OIP3 value for the approximately 3.5dB insertion loss of the filter. IIP3 will not change because the RF input to the mixer has not changed. OIP3 will because it is an output referred value. Next, we can attenuate the mixer IF output by 20dB to prevent overloading the spectrum analyzer’s front-end mixer and improve the impedance match. 20dB is used arbitrarily and a significant enough attenuation value should otherwise be used. 22. IF output spectrum of the MM1-0312HS with the block diagram setup from figure 18 with a DPX-0R5 lowpass filtering the IF output and a 20dB attenuator. If we add 20dB back to the above OIP3 measurement, we see no negative impact to the measurement; just an additional hoop in the IP3 calculation. So we can disregard this source of error unless the measurement calls for it. An example of when this could be necessary would be when the DUT (device under test) mixer’s IF output has a high power (>+5dBm) or when the spectrum analyzer’s internal attenuation is insufficient. Error Source #5 From a mixer use standpoint, the argument in Error Source #4 makes sense. To prove it, what if we remove all attenuation into the spectrum analyzer? 23. Measurement from figure 21 repeated with 0dB internal attenuation on the spectrum analyzer. The remeasured OIP3 is +15dBm. To interpret the result, do we say that the measurement improved or is this an erroneous result? We can verify this by removing the DUT. If we compare measurements of the OIP3 of figure 21 with and without internal attenuation turned on in the spectrum analyzer, we see that the measurement significantly degrades. 24. Spectrum of the 2-tone RF input into the MM1-0312HS with the block diagram from figure 20. 20dB internal attenuation in the spectrum analyzer. Identical to figure 19. 25. Spectrum of the 2-tone RF input into the MM1-0312HS with the block diagram from figure 19. 0dB internal attenuation in the spectrum analyzer. Because the measurement of the 2-tone RF input without attenuation is noticeably different than the measurement without attenuation, it’s very likely the spectrum analyzer’s front end mixer was overloaded and gave an erroneous result when no attenuation is applied. This is the same case for the MM1-0312HS and a measurement without attenuation protecting front-end mixer of the spectrum analyzer should be disregarded. There should always be some attenuation present to protect the front end mixer of the spectrum analyzer from being overloaded. Typically, 10-20dB attenuation is sufficient. In addition, it is always good practice for the instrument to be run in a as low a resolution bandwidth (RBW) as possible. The reduced resolution bandwidth allows for a slower, more precise measurement to be taken. This reduces the measurement bandwidth of the analog to digital converter within the spectrum analyzer. A resolution bandwidth of 10Hz will measure a tone with 10Hz instantaneous bandwidth versus a resolution bandwidth of 300KHz which will attempt to measure a tone with up to 300KHz instantaneous bandwidth. Using a slower resolution bandwidth avoids measuring the side of the tone aswell as improving the noise floor of the measurement. Generally, the resolution bandwidth will automatically adjust and lower itself when the frequency span of the instrument is set. It can also be manually set for most modern test equipment. Refer to the operator manual for guidance. Testing Part 2: Final Test 26. Final block diagram of the recommended mixer IP3 test setup on a spectrum analyzer. Optional blocks are highlighted. The block diagram is the final test setup in figure 29. Its measured MM1-0312HS IP3 data is below in figure 29. For the measurement in figure 30, the LO filter was omitted. Practically, the high order spurs that mix off the 2nd LO input harmonic that fall on top of the $(2f_1 - f_2) - f_{LO}$ and $(2f_2 - f_1) - f_{LO}$ intermodulation spurs should be at a much lower power than the intermodulation products. Like the results from section Error Source #4, the LO filter is a test condition dependent block. In particular, it is undesirable to filter the 3rd harmonic if you’re driving a classic, hybrid T3 style mixer with a square wave LO; less so if you’re using a newer MMIC T3. The reason behind this is, because for a T3 style mixer, the 3rd and 5th harmonics of a square wave LO remixes and biases the diodes within the mixers in such a way that the mixer nonlinearity is improved over a normal sine wave LO. For a double balanced mixer, the LO filter may or may not be necessary. Typically, it is not because the LO’s second harmonic can be pushed out of band of the mixer. If we retest the MM1-0312HS with the block diagram from figure 28, we should see an improvement compared to the block diagram from figure 5. 27. IF output spectrum of the MM1-0312HS with the block diagram setup from figure 26. Compared to the quick and dirty measurement setup and mixer IP3 data: 28. IF output spectrum of the MM1-0312HS with the block diagram setup from figure 5. Identical to plot to figure 6. There was an improvement of 0.5dBm in the measurement of the mixer’s IP3. This is only slightly more in-line with the VNA measurement. This 0.5dBm change could practically be only attributed to measurement uncertainty. The measurement uncertainty in our Rohde and Schwarz FSW85 is 0.3dB for this measurement frequency. Practically speaking, the IP3 measurement between both test setups are virtually identical. So confidence in the accuracy of the measure can be confirmed. However, the real improvement is in the IP3 measurement of the 2 tone RF input. 29. Spectrum of the 2-tone RF input into the MM1-0312HS with the block diagram from figure 5. Identical to figure 7. Final Test Setup: 30. Spectrum of the 2-tone RF input into the MM1-0312HS with the block diagram from figure 18. Identical to figure 19. What this means is that we should be able measure very high IP3 values because the measurement is closer to ideal conditions where the input intermodulation products are not present and mixing down to the $(2f_{RF_1} - f_{RF_2}) - f_{LO}$ and $(2f_{RF_2} - f_{RF_1}) - f_{LO}$ frequencies. Choosing the MT3-0113LCQG,and instead of a +15dBm LO, we use a higher power +25dBm square wave LO and repeat the comparison. The higher LO power should give a high IP3 measurement and push the measurement system to its dynamic range limit. The test is: Will the improvement to the 2-tone RF input IP3 allow for high mixer IP3 measurements? Using the block diagram from figure 5, we compare the quick and dirty mixer IP3, taken with the newly specified +25dBm square wave LO: 31. IF output spectrum of the MT3-0113LCQG using the block diagram from figure 5 with a +25dBm square wave LO. and final test setup IP3 measurements: 32. IF output spectrum of the MT3-0113LCQG using the block diagram from figure 26 with a +25dBm square wave LO. There is an improvement of 1.5dB in the measured mixer IP3. This is a 40% improvement in the measured value over the quick and dirty mixer measurement. If the same test setup is used but the mixer device under test from the MT3-0113LCQG to a T3-03MQP or any other mixer, what can be seen is that the dynamic range of the spectrum analyzer was reached or very close to being reached. While the mixer IP3 measurement has improved, there is no such improvement that can be made without using a better spectrum analyzer. If we test the H diode version of the mixer which differs from the L diode version of the mixer based on the forward turn on voltage of the mixer’s diodes, the MT3-0113HCQG, we see that the resulting measurement is the same. 33. IF output spectrum of the MT3-0113HCQG using the block diagram from figure 26 with a +25dBm square wave LO. What can be inferred from the H diode’s measurement is that for this IF frequency ~300MHz, the spectrum analyzer can only measure the up to a ~+25dBm OIP3. The spectrum analyzer hit its dynamic range limit. It is an expectation that a higher barrier diode variant of the same mixer (same circuit typology, RF/LO baluns, wiring, etc.) to have better linearity than its lower barrier version. The higher barrier diode version of a mixer should have a an IP3 of $20\log{\frac{V_{F_H}}{V_{F_L}}}$dB higher than the low barrier diode mixer; where $V_{F_H}$ is the forward turn on voltage of the high barrier diode and $V_{F_L}$is the forward turn on voltage of the low barrier diode. This can be verified against the datasheet. For further comparison for the argument of which diode variant should be superior, measurement of identical mixers with 2 diode variants such as the MM1-0626HSM and MM1-0626SSM should be explored. Both MT3 variants’ corresponding datasheets agree that for the same test conditions on a VNA, the measurement of IP3 at 6GHz RF differs by approximately 3dB; the datasheets can be found here for the H diode version and here for the L diode versions. For a T3 style mixer, the difference in diode version matters to a lesser extent, a similarly strongly driven L and M diode T3-03, for instance, will typically have a very similar IP3 value. This also holds true for the MT3s. To improve a mixer’s IP3, circuit topology in addition to diode forward turn on voltage matters. Conclusion A +5dBm improvement in the 2 tone RF input IP3 is a lot. It’s a 10dBc improvement in the IMD3 spur suppression relative to the IF down converted fundamental frequency. For very specific circumstances, where the front-end mixer of the spectrum analyzer is in a “sweet spot” this improvement can greatly improve an IP3 measurement. The time and capital consuming test setup for high IP3 measurements can be completely unnecessary. The ability to do the measurement depends highly on the dynamic range of the 3 synthesizers and the spectrum analyzers used. For the measurement provided in this document, it appears the front-end mixer of the spectrum analyzer is the limiting factor stopping the measurement of mixer IP3 values higher than ~+31dBm. This is evidenced by the decreased measurement resolution when spectrum analyzer input attenuation is removed and the disagreement between the measurement of the high barrier diode mixer and with its sister low barrier diode variant. For a normal, or even good, mixer measuring the IP3 using a quick and dirty setup is not punishing. Results agreeing to within 1.5dB against the published datasheet is very good considering the time and capital costs (approximately$150K USD) of setting up a proper IP3 measurement station. Results are very much acceptable using a bare-bones test setup. In order for any of the solutions in this document to be of use, the dynamic range of the measurement equipment must be higher than what was used to gather the presented data and the DUT mixer must have an extremely high IP3.

The RF/Microwave parts used for the test station can be found here for high isolation power dividers and here for high suppression filters. Look forward for high >+35dBm IIP3 highly linear MMIC T3 mixers covering high frequency bands in the near future. Contact info@markimicrowave.com for more information regarding all products.

## GaAs or GaN: Best Choice for Highly Linear Mixers?

Everywhere you look in the microwave industry press or at the international microwave symposium you see one topic mentioned over and over again: Gallium Nitride (GaN). There is presently a gold rush in the industry to produce new varieties of products in GaN. The wide bandgap and high electron mobility of GaN mean that it is capable of a much higher power density than Gallium Arsenide (GaAs).  The introduction and availability of a 0.15 micron commercial GaN processes means that fabless integrated circuit companies (such as Marki) can produce GaN microwave products at frequencies comparable to GaAs. This has led to a profusion of products including power amplifiers, low noise amplifiers, driver amplifiers, and many other amplifiers using GaN that set new records for power at high frequency.

One thing that is fueling the drive to GaN is the increasingly stringent linearity requirements placed on modern RF/Microwave systems. Since the noise floor of a system can only be reduced so much, the only way left to increase the dynamic range is to increase the power compression level and decrease the intermodulation products to create a wider spur free dynamic range. At the heart of most RF receivers is a mixer that limits the linearity and therefore the dynamic range of the receiver. So the next step looks obvious; GaN is increasing the dynamic range of many components in the system, so it should increase the dynamic range of the mixer as well!

There are five main reasons that this logic doesn’t work:

## MM1-2567 Operation to 80 GHz

We occasionally receive requests for a mixer that will operate above our highest frequency mixer, the MM1-2567LS. The truth is that, contrary to the datasheet, the MM1-2567 actually operates above 67 GHz.  It was designed to operate up to a frequency of 80 GHz on the RF and LO sides. The issue is that our test equipment can only measure conversion losses up to 67 GHz directly. To solve this problem, I devised an experiment to prove whether the mixer frequency actually extended to it’s simulation limit or not. Here is the experimental setup:

## Repeatability of T3 Mixers and Other Handmade Microwave Components in Six Charts

T3 mixers are the highest dynamic range mixer available. They are also handbuilt parts, subject to unit to unit and lot to lot variability. In this blog post we attempt to quantify that variability. Our sample is 10 T3-08LQP mixers from 5 different date codes. All the date codes are separated by at least a month, totaling nearly two years.  Therefore, the variation you see in the plots below accurately represent the variation a designer could expect across two years in the life of their product. Of course there are always outliers, but the following represents typical performance variation.

Conversion Loss

## New MMIC Mixers from Marki Microwave Cover 3-24 GHz

Marki is bringing advanced mixer designs to a broader market with four new models of GaAs Schottky diode double balanced mixers covering S and K band applications.  These designs combine the legendary mixer design expertise of Marki Microwave with the repeatability and economies of scale intrinsic in the MMIC production method.

## What happens when you underdrive a mixer?

The oldest question in mixer tech support is probably “what happens when I drive the mixer with X dBm LO?”, where X is some number lower than what we recommend. In general, and particularly in the past, we have avoided this question. Our recommendation was and is to never underdrive a mixer. The reason for this is that a mixer with insufficient LO drive does not act as a switching device, but as a square law device. If the LO does not turn on the diodes then the physics of mixer operation change completely, and all of our carefully laid design work is thrown out the window.

Indeed, when you underdrive a mixer the conversion loss is not the only thing that changes. All of the specs change, and in unpredictable ways. The LO side in particular responds weirdly, because a lot of deficiencies and inefficiencies on the LO side are exposed when it is underdriven and concealed under normal operation. In this post we will show all the bad things that can happen when you don’t supply an adequate LO drive to the mixer, and then leave it to you as the user to decide what LO drive to design with.

## Microlithic Die Attach

We are often asked what the ‘preferred’ method is for attaching Microlithic die to the substrate is. We use, and prefer, silver epoxy die attachment, because epoxy die attach

• is a low temperature process (epoxy can dry at room temperature, vs. very high temperatures for solder or eutectic)
• is easy to do, with a low learning curve and minimal waste of parts
• does not require a solderable die surface (not all chip based Microlithics have a nickel solder barrier layer)
• has a low probability for voids underneath the chip
• is a reversible process, where the part can be removed without damage if performed carefully.

For these reasons, we use silver epoxy (specifically Epotek H20E), place a minimal amount beneath the die, and place the chip into position so that a thin epoxy fillet is created around the chip. The part is then cured at temperature according to the epoxy instructions.

## All About Mixers as Phase Modulators

In our last post we showed the physical basis for how mixers are used as phase detectors, concluding by showing that IQ mixers make ideal phase detectors due to their ability to unambiguously identify the relative phase between two signals at any power level. In this post we examine the opposite: how to use mixers as phase modulators. It seems like you should be able to use them in exactly the opposite way, which is to apply a DC voltage to get a linear phase shift. Unfortunately, it’s not that simple.

Note: as with mixers as phase detectors, we as the manufacturers are not the best experts, but our users are. In this case I would recommend Kratos General Microwave, whose application notes I used in preparation of this blog post.

Why phase modulators?

Before examining how to get a phase modulator, let’s look at why you might need one. The main applications are communications and electronic warfare.

Communications: Phase modulation (mathematically identical to frequency modulation) has been used since very early in radio communications, due to FM communications having constant amplitude, better spectral/power efficiency, and convenience. The most common way of understanding phase modulation is with binary phase shift keying (BPSK), or quadrature phase shift keying (QPSK) if both orthogonal components are used. All modern communication systems use these techniques, so they have been written about very extensively, and we will assume that you are familiar with them.

Electronic Warfare: Here it gets interesting. If you have a phase modulator in a jammer, you can trick an enemy radar system into thinking that your plane/boat/tank is not where it actually is. You do this by listening to their radar pulses and responding with frequency shifted radar pulses, making it appear that you are moving at a different speed. This is the classic decoy technique. Modern jammer systems employ much more advanced, exotic, and classified schemes than this that I hope I never have the classification level to learn about. The principles, however, are the same.

Double Balanced Mixers as Phase Modulators

Let us start by running a phase detector in reverse. Instead of a DC output, lets input a DC signal to the IF port and a CW signal to the LO port and see what comes out of the RF port. If the device is reciprocal, then a small DC voltage/current should induce a small phase change, and a larger voltage should create a larger phase change. Here is what happens:

So this is nothing like what we expected. Why? I don’t really know. Somewhere the hand waving ‘superposition’ argument I gave in the last blog post breaks down, and something is not reciprocal. The above behavior makes sense from what we know about double balanced mixers. Namely that with no voltage applied, the LO-RF isolation prevents any signal from passing through. As the DC voltage is applied, it breaks down the symmetry of the diode quad, reducing the isolation and allowing more of a signal to pass through, although without a phase change. Now here is what happens when we apply a negative voltage:

So at least with a negative voltage you can get a 180° phase flip. So a double balanced mixer does give you phase modulation, but only between two different options. This makes it suitable as a BPSK modulator (emphasis on the word binary), but not for much else.

IQ Mixers as Phase Modulators

IQ mixers worked great as phase detectors, will they work great as phase modulators? Only one way to find out:

This picture is a little confusing, but the idea is that it works out mostly okay. As with the double balanced mixer, you see no signal pass through with 0 voltage applied. If you manipulate the DC voltage applied to the I and Q port, then you will see the phase rotate around the complete circle as expected. When you only have one voltage applied you will be almost at 0°, 90°, 180°, or 270°. There are some phase errors, I would imagine due to the non-ideality of the components, but this can probably be accounted for. There is also a non-uniformity to the amplitude due to the isolation being degraded differently in the different mixers.

So what is going on here, how can you achieve an arbitrary phase with an IQ mixer when you can only get two phases with the double balanced mixers that make them up? This is the following trigonometric identity:

a cos(x) + b sin(x) = R cos(x-theta)

where R2=a2 + b2 and tan(theta) = b/a. This means that by simply changing the values of the input signals (in this case by modulating the isolation of the mixers) you can achieve any phase within the range of the tangent function (-90° to 90°), and then by flipping the value to negative you can achieve any value in the phase circle.

Is this a good idea?

Just because you can do something doesn’t mean you should, you can drive a car with your feet if you want to but that doesn’t make it a good idea.

The short answer is yes and no. Using an IQ mixer is an easy way to achieve an arbitrary phase modulator, definitely useful if you are in the lab. A few problems though:

• The isolation change with applied DC voltage is non-linear, and the whole structure has to be carefully characterized to achieve repeatable results.
• All non-idealities are frequency dependent, so this characterization has to take place at each point in the system. Further, if you are modulating a broadband or multitone signal you won’t be able to correct for the system errors at all frequencies.
• The insertion loss when used in this way will look very little like the insertion loss shown in the datasheet for the IQ mixer. For example, here is the MLIQ-0416 datasheet conversion loss:

Nice and flat across the band around 8 dB. Now here is the same mixer as a phase modulator:

This starts out strong, but then falls off at higher frequencies. This is because the second plot is of an insertion loss instead of a conversion loss. The difference is subtle but important. In the first case the LO is only used to turn on the diodes, so the losses that it takes passing through the LO quad hybrid don’t matter very much. In the second case, the LO is the signal, so any loss that it takes shows up as an insertion loss.

Those are the main problems, and if these can be overcome, then you may be in business.

Single Sideband Mixer as a Phase/Frequency Modulator

There is another way to use an IQ mixer as a phase or frequency modulator, and this involves creating quadrature CW signals into the IF port of the mixer, effectively using the IF port as the LO, and varying the frequency of this LO signal to change the frequency offset. We will examine this, and other use cases, in our next blog post on using IQ mixers as single sideband upconverters and image reject downconverters.

## All About Mixers as Phase Detectors

Some of the most common questions we receive here are about using mixers as phase detectors. We previously discussed this topic in the post, “DC Offset and Mixers as Microwave Phase Detectors”. In this post we will go into much further depth about the physical mechanisms by which mixers act as phase detectors, and what is important for engineers trying to accomplish this in the lab. First a warning though: we’re just showing experimental results here. The real experts in phase detectors, phase noise, and all things related to phase are the people that do this every day at Holzworth Instrumentation.

Double Balanced Mixers as Phase Detectors

Much has been written about how double balanced mixers work as phase detectors (for example, see this article from Watkins Johnson about the subject). As with most circuit topics the descriptions in the literature are based in math rather than physical principles, so we’ll now consider the physical mechanisms in play when a double balanced mixer is used as a phase detector. Let’s look at what happens when we apply two in phase (frequency matched) voltage signals to an ideal double balanced ring mixer1:

The phases correspond to the phase of each signal as it appears at the diodes. Only two are show, but take my word that superposition works here and every other in between state produces the same effect. For current to flow, two conditions must be met. First there must be a voltage differential across the diode. Second is that the diode has to be pointed in the correct direction. The red arrows indicate where those conditions are met and current will flow. As you can see, when the signals are in-phase current will flow into the IF balun, creating a positive DC voltage at the IF port. Now out of phase signals:

The situation is similar for out of phase signals, except that current is always pulled out of the IF balun, thus creating a negative DC voltage at the IF port. For quadrature signals, there is equal current flowing both into and out of the IF balun. This means that no net DC current is created, no net voltage is apparent. The IF port is essentially always a DC virtual ground. This is the physical basis for why a double balanced mixer will show no DC voltage for two signals in quadrature2.

With these principles understood, let’s go in the lab and see what happens when we actually apply these signals to double balanced mixers. First we create two signals and use an oscilloscope to verify that they are in quadrature:

At 10 GHz the period is 100 ps, so 25 ps out of phase is in quadrature.  Now we apply these voltages to the input of a ML1-0220LS mixer, and what do we find? Nonzero voltage! In fact, here is what the DC output voltage (taken with a terminated bias tee, this is very important) looks like a function of phase between the two input signals:

Now we would expect this to be a peak at 0°, and the minimum to be at 180°. What is going on? This is a phenomenon that is documented by Stephan Kurtz in the previously referenced WJ app note. In modern double balanced mixers the RF and LO baluns are not identical. In fact, they are not even close. One side is built as a magic tee, where the IF is removed, and the other side has a return to ground on it. Even though the LO and RF baluns traditionally cover identical frequency bands, there is no reason that they need to. They can be completely different! This means that they most likely have a different electrical transmission length and phase delay, which is why the peak of the voltage curve is not quite at 0°. Another effect highlighted in this app note is that there is a voltage offset that shifts the entire curve up (or down). As we detailed in our first post, the excellent balance and isolation of the ML1-0220 minimizes this DC offset and makes it not noticeable for this plot.

1 Note that the necessary DC current return to ground path is necessary but not illustrated for clarity.

2 It is easy to imagine how to extend these principles to the situation where the signals are not at their peak or zero values, and similarly to phases that are not either perfectly in phase or out of phase. While superposition does not strictly work in a nonlinear system such as this, the results one would expect from superposition are maintained qualitatively.

IQ Mixers as Phase Detectors

Now we can calculate the phase of the signal. Excellent. However, there are two ambiguities that we need to clear up. Since this is a sine wave instead of a sawtooth wave, there is some ambiguity about the phase. The same output voltage could be two different phases, except for the max and min. This is fine if you are doing phase noise testing, where you put the two signals in quadrature and just look at any voltage that comes out. For actually detecting the phase between two signals though, it isn’t enough information. The second ambiguity is that we need to know the max and min voltage levels, as well as the DC offset, to determine the phase. Since the DC offset in Microlithic mixers is small we can ignore this, but we still have a problem if the incoming signals change power at all.

How do we resolve this? One way is to use two mixers as phase detectors and deliberately introduce a phase shift between the two inputs (RF and LO). 180° is no good, because the phase ambiguity remains, so a balun is out. A length of line changes phase with frequency, so that is out too. The other broadband phase shifting options we have are a Schiffman phase shifter or a quadrature hybrid. The quad hybrid is much more common and easy to build3, so what would a structure with a quadrature hybrid introduced on one side look like?

That’s right: the structure is exactly an IQ mixer. Since I and Q are in quadrature, it is easy to calculate the phase between the two signals as

after making a small correction to scale the I and Q values by their peak output level and DC offset4. Let’s look at the same plot of voltage vs. phase for the IQ mixer, along with the calculated phase:

As expected, the calculated phase is almost linear with input phase after the correction factors. This is a significant improvement over the double balanced mixer, since we don’t need to know the input power levels and there is never any phase ambiguity. But how close is the IQ mixer to invariant with input power? When the two signals are at 0/90/180/270 degrees to each other, there is obviously very little variation in calculated phase with power since one of the voltages doesn’t change. If we pick a phase in the middle (135°)5, this is what it looks like:

As you can see the power levels the agreement to 135° is excellent. As we increase to higher power levels, one of the mixers compresses sooner than the other mixer, and the phase is thrown off. Up to 0 dBm, however, the agreement with the real phase is excellent. This does not address what happens when one of the signals is significantly higher than the other one, nor with double balanced mixers when you are just trying to detect phase changes, where high powers are desirable to increase sensitivity.

Now that we have examined the physical mechanisms of how mixers work as phase detectors, we can do the reverse and see how they work as phase modulators. This is the subject we will tackle in our next post, “All About Mixers as Phase Modulators”.

3Quad hybrids are easier than Schiffman phase shifters, but still ridiculously difficult to build broadband. You don’t have to trust me, you can try yourself, and then buy ours when it takes you 6 months.

4You also have to convert to a -180 to +180 phase range, or 0 to 360, or whatever. Arctangent only gives you values from -90 to +90, so you have to use the sign of the signals to figure out where exactly you are.

5How do we know the phase is 45 degrees? Because we put the signals in quadrature (which is the same at any power levels), and then moved them 12.5 ps on the oscilloscope, equivalent to 45° at 10 GHz.

## The T3: A High Dynamic Range Mixer for 4G, LTE, and 5G testing

Spectral regrowth is a big deal for you. In order for the wireless revolution to continue apace, enabling you to watch funny cat videos faster in more crowded environments, spectral regrowth must be conquered wherever it occurs. Spectral regrowth is what occurs when a broadband or spread spectrum signal intermodulates with itself, creating deterministic products that look like noise, limiting the signal to noise ratio of the signal. According to Shannon’s theorem this limits the information capacity of the signal, and thus your video takes longer to load (for some reason this always happens at the worst time).

Spectral regrowth comes from a handful of sources. It can come from mixers, but in installed communication systems it tends to come from the power amplifiers at the transmitter and the connections to the antenna itself (called ‘passive intermodulation’ or PIM). It is made much worse by using higher power and by denser concentrations of signals. Both of these factors are increasingly common as data capacity is increased. This is why highly linear amplifiers and PIM are both big buzzwords in the mobile communications world right now.

What is not always talked about is that ‘spectral regrowth’ in the mobile communication world is the same as ‘two-tone intermodulation distortion’ or ‘IP3’ in the microwave world. Two tone modulation distortion is what causes spectral regrowth, just summed over all of the frequencies involved in the signal. This can be seen by moving from two tone testing to three tone testing in a standard double balanced mixer, the Marki M1-0212SA:

As you can see, it is a complete mess! This is considered very good for a double balanced mixer, and it is better than you would get from any GaAs mixer on the market. This is with 0 dBm output signals and a 25 dBm LO drive (square wave, this will matter in a minute). Note that we use 0 dBm output power as the reference instead of the input power. As we have mentioned before, this is what you really care about as a designer (how much range do I have at the final stage in my receiver) and accounts for the variation in insertion loss between mixers. You can cheat by adding loss to a mixer to improve the input IP3, but you can’t cheat on the output IP3.

This output spectrum is obviously unsuitable for operating with, nonetheless testing a high performance system. The testing system must have at least 20 dB more dynamic range than the system itself, and usually much more, so getting rid of these obnoxious intermodulation products is critical. This was the state of the art for many years, then Watkins Johnson came out with their termination insensitive mixer, the M8TH (still on sale from Ma/com and still considered the standard by some). Here is the M8TH output spectra:

Much better!  The intermodulation products have been suppressed significantly, although they are still significantly limiting the dynamic range to 45 dB. Recently there has been much talk of the FET mixer. At Marki we have not been that impressed with the FET mixer, because of the narrow bandwidth over which it works. Combined with the intrinsic poor isolation available in the FET circuit, this limits the performance below what we like to see. Nonetheless people get excited about it, so here is the narrow bandwidth, poor isolation PE4140 FET quad mixer we built two and three tone results:

Good improvement above the termination insensitive mixer, now the dynamic range is up to 50 dB. At least I guess it is cheap and good for communication applications, but with the narrow bandwidth and isolation problems it isn’t well suited as a test and measurement mixer. Now let’s look at the T3. As a true commutating mixer it is in its element with a 25 dBm square drive. Here are the results:

WOW. Another 10 dB+ of two tone improvement, above the ‘state-of-the-art’, for a 60 dB dynamic range. Here they are together for comparison:

And this is also the mixer with the best spurious suppression, and isolations and conversion loss as good as any mixer available. All of this across a 2000:1 bandwidth ratio. It isn’t fair to say that the T3 is the Ferrari of mixers. It’s more like the T3 is a fighter jet racing a car. When it comes to choosing a mixer for your test system, there really is no choice but the T3.